Voltage down converter for high speed memory

ABSTRACT

A voltage down converter (VDC) applicable to high-speed memory devices. The VDC includes a steady driver and active driver along with at least one additional transistor. The steady driver and active driver are coupled by a transistor switch during device start-up to provide fast ramp-up to operating voltage and current. After start-up, the steady driver and active drive function to maintain a steady operating voltage and current. An additional transistor is digitally controlled to drive up operating voltage and current upon issuance of an active command representing read, write, and/or refresh of memory. In this manner, the additional transistor provides fast compensation for fluctuations in operating voltage and current brought on by activity in the memory array.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a divisional of application Ser. No. 11/781,581,filed Jul. 23, 2007, which is a continuation of application Ser. No.11/195,641, filed Aug. 3, 2005, now issued as U.S. Pat. No. 7,248,531.

FIELD OF THE INVENTION

The invention relates generally to a voltage down converter thatprovides a supply voltage and current to a memory device to minimizevoltage supply variation and improve power consumption. Moreparticularly, the invention relates to a voltage down converter used inhigh speed memory devices.

BACKGROUND OF THE INVENTION

Voltage Down Converters (VDCs) are used to lower the level of anexternal power supply voltage (e.g., Vddq) provided to a semiconductordevice to a desired internal power supply voltage (e.g., Vdd). Forexample, a voltage down converter in a semiconductor integrated circuitmay lower an external power supply voltage to the level of an internalpower supply voltage, so that each component element within theintegrated circuit may be operated with the internal power supplyvoltage to reduce power consumption and secure sufficient reliability ofeach component element. Another use of the VDC in the area of high-speedmemory devices such as dynamic random access memories (DRAM), is thegeneration of a substantially constant high supply voltage that isgreater than the logic “1” voltage level stored in a memory cell.

A fast voltage down converter can be used as a substantially constanthigh voltage supply for providing a power supply voltage (e.g., Vpp) tothe memory device. This is particularly important within high speedembedded memory devices having small die area available on asystem-on-a-chip (SOC) application. When the die area is small, i.e. inan embedded memory implementation, there is inadequate capacitiveloading on a high voltage supply output in order to store the chargenecessary to maintain a substantially constant supply voltage. Moreover,the operating frequency of current memory devices now exceeds 800 MHz inmost leading edge embedded memory applications. A fast response of thevoltage down converter is an important factor in providing asubstantially constant high supply voltage Vpp to the memory. Thestability of the substantially constant high supply voltage Vpp duringthe different modes of operation of the memory, greatly depends on thecapacitive loading and the reservoir capacitance of the voltage supplyoutput. The capacitive loading is created by the sum of all elementshaving an inherent capacitance (for example transistor gates) that aremutually connected to the Vpp output node.

The reservoir capacitance is typically a separate dedicated largecapacitance that is connected to the Vpp output to provide furthercapacitive loading and stability on the output in addition to theinherent aforementioned capacitive loading on the Vpp output. However,the small size of embedded memory devices creates some problems ofstability in the required internal level supply. Active operations thattake place in the memory, i.e. read, write and/or refresh operations,contribute to create a ripple effect on the voltage level Vpp given thecurrent load that such operations provide. The voltage ripple level isaffected by the total capacitance such that

V_(rippLe)˜I_(LOAD)*T/(C_(LOAD)+C_(RESERVOIR)),

where ILOAD is the leakage or operating current and CLOAD is the totalloading of the Vpp voltage. Since the load and reservoir capacitancesare small in embedded memory applications, the ripple voltage levelbecomes very dependent on fluctuations in load current, which can occurbased on different current loading scenarios. Because of this problem,embedded memories experience significant voltage supply fluctuation.

The common solution for this fluctuation problem has been to insert aslarge a reservoir capacitor as possible in order to hold the voltagelevel provided by the voltage down-converter VDC as stable as possible.One such conventional solution 100 is shown in FIG. 1A. However, a keyissue of embedded memory is their small size that does not allowadequate space in order to place such a large reservoir capacitor. Insuch instance as in FIG. 1A, either CLOAD or CM would have to beadjusted to be a very high capacitance in order to provide the necessarystability for the voltage-down conversion. Because CLOAD represents thewhole memory array and such arrays are becoming increasingly smaller,the result is that CM would need to be increased significantly. However,increasing CM results in an increased penalty area that, in the case ofembedded applications, is one of the major design constraints.

An alternative approach as shown in a prior art voltage down converterFIG. 1B is to add an additional transistor to the output of the voltagedown converter in order to supply fluctuating current demands of thememory. However, this solution is not appropriate for embedded memoriesthat have many diverse operations such as single read and write or reador write concurrent with a refresh operation, each type of operationsinking different amounts of current depending on the process, voltage,or temperature variations. Furthermore, the inherent delay of thecomparators used in this approach is not sufficient to maintain a steadyvoltage level in the voltage down converter output. This is especiallythe situation in memories operating at high frequencies.

Several other conventional VDCs are shown in U.S. Pat. No. 6,806,692issued to Lee. In such memory devices, a VDC may be used in variousapplications to supply large amount of “AC” type current (e.g.,alternating or fluctuating current) while sustaining the necessary “DC”type current (e.g., a fixed current) and a steady DC voltage level. Theconventional VDC design has some drawbacks when employed in modernapplications, which often require large, fluctuating output currents. Toprovide a large current, the VDC will typically require a relativelylarge source follower transistor. In order to drive such a largetransistor, a corresponding comparator has to be relatively powerful.The relatively large transistor and comparator size results insubstantial and undesirable current consumption for the VDC. Also, thefeedback or coupling capacitor must be relatively large in size tostabilize the VDC. The performance-to-power ratio of this type ofconventional VDC diminishes with increasing current supply requirements.In the presence of increasing current demands and high frequencies ofoperation, the conventional VDC eventually becomes sluggish to supplyadequate AC current for digital circuits. When such a scenario occurs,the VDC cannot maintain the output voltage Vpp at a steady level andvoltage level dipping occurs. In a DRAM chip, a large dip in voltagelevel can cause memory cells to fail. Accordingly, there is a need tokeep Vpp stable even while the memory device is operated at highfrequencies for various operations each requiring high current demands.

The VDC 110 in prior art FIG. 1B includes comparators 170 and 171, arelatively large PMOS (positive channel metal oxide semiconductor)transistor 151 and a relatively small PMOS transistor 152, and afeedback compensation or coupling capacitor Pcc. Comparators 170 and 171receive two inputs, a reference voltage (Vref) input and feedback loopinput (Vfb). The output of the comparator 170 is coupled to the gate oftransistor 151 while the output of the comparator 171 is coupled to thegate of transistor 171 and to the gate of transistor 153. Transistor 170has a high gain but a low speed response to the signal that it receivesin its gate, while transistor 171 has a low gain but a high frequencyresponse.

The prior art VDC 110 as shown in FIG. 1B does not operate well for highfrequency memories. This is due to the fact that the analog circuitshown by VDC 110 is inherently slow and cannot function well within everincreasing memory speeds where activity on the load is quickly changing.More specifically, the comparator requires a minimum difference betweenthe reference voltage and the feedback voltage in order to change itsoutput and to increase or decrease the current provided to the outputnode D1.

There is therefore a need for a new and improved voltage down converterfor use with high speed memory devices, which can provide relativelylarge output currents and voltages, which minimizes voltage variationsduring operation, and which has improved stability and robustness.

SUMMARY OF THE INVENTION

An object of the invention is to remedy the drawbacks set out above byproviding a voltage down converter that includes an active driver havinglittle driving capability with additional transistor pairs havingseparate digital control outputs and precisely controlling thedown-voltage and stabilizing the down voltage level without adding ahuge reservoir capacitor.

In an embodiment, the present invention includes a method for supplyinga voltage to a random access memory circuit within an integrated circuitdevice, the integrated circuit device having an input pin for receivingan external high voltage, the method including connecting the input pinthrough a series pass transistor to a power supply voltage output nodeand regulating said series pass transistor to generate a substantiallyconstant high power supply voltage that is greater than a logic “1”voltage level stored in a memory cell.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present invention will now be described, by way ofexample only, with reference to the attached Figures, wherein:

FIGS. 1A and 1B are examples of prior art voltage down converters;

FIG. 2 is a schematic diagram of a voltage down converter in accordancewith an embodiment of the present invention;

FIG. 3 is a graphical representation of power-up, standby mode, andconsecutive active mode in accordance with the present invention;

FIG. 4 is a simplified version of the schematic diagram of FIG. 2showing features of power-up and standby modes in accordance with anembodiment of the present invention;

FIG. 5 is a simplified version of the schematic diagram of FIG. 2showing features of gate control of additional transistors provided inaccordance with an embodiment of the present invention; and

FIG. 6 is a graphical comparison of the schematic diagram of FIG. 2 inaccordance with an embodiment of the present invention relative tooperating voltages without benefit of the present invention.

DETAILED DESCRIPTION

With reference to FIG. 2, a Voltage Down Converter (VDC) 200 inaccordance with one embodiment of the present invention is shown. TheVDC of the invention is designed to generate a substantially constanthigh voltage VPP, that is greater than the logic “1” voltage levelstored in a memory cell. The present VDC uses the external high voltageprovided to an input pin of an integrated circuit that has a memorycircuit, in order to generate the VPP voltage without the use of acharge pump. The VDC includes a standby main driver 211 with high gainand low frequency response, an active driver 212 with low gain and highfrequency response, and two additional transistors 213, 214, and therelevant circuitry such as a regulator 230, 231. The main driver 211 andthe active driver 212 are implemented in a series pass configuration.The two additional transistors 213, 214 are in the form of PMOS driversand receive in their gate a digital control signal. For clarity ofillustration and for the purposes of distinguishing such additionaltransistors from the standby 211 and active drivers 212 mentioned withinthis description, such additional transistors 213, 214 will behereinafter referenced as “active charge injector transistors.” However,it should be well understood that such nomenclature including the term“charge injector” should not be considered as limiting such elements toany previously understood or known definition of “charge injectortransistor” or “active charge injector transistor.”

The charge injector transistors 213, 214 are used to provide a pulsedincrease in the current into the VPP voltage power supply node, tocompensate for the load current increases occurring during read, writeand/or refresh active operations. The charge injector transistor isimplemented in a series pass configuration. The signal that controls theoperation of the charge injector transistors 213, 214 is generatedduring the read, write and/or refresh operations or in anticipation ofsuch operations. Preferably, the dimensions of the charge injectortransistors 213, 214 are designed to provide the necessary current thatis sufficient to compensate for the increase in the load during anactive cycle. The time that the control signal is applied to turn on thecharge injector transistors is set so that the required amount ofload-compensating current is supplied through these transistors 213, 214to the VPP voltage output, so the VPP voltage level is maintained stableduring the read, write and/or refresh operations. In a furtherembodiment, the time of the active pulse is programmed to compensate fortemperature, voltage and process variations.

FIG. 2 includes a regulator that is shown as two comparators 230, 231that include a PMOS current mirror load and two NMOS (negative channelmetal oxide semiconductor) series-connected differential inputs and biascontrol NMOS connected to ground. Decoding logic is used to combine theread, write, and refresh operations with information of which single ordouble memory bank is accessed, to generate a control signal to enableone active charge injector transistor. Such decoding logic 240, 250, 260and level shifter elements 251, 261 are embodied by appropriatecircuitry (e.g., D-flip flops and logic gates) well known within thearea of digital circuits and need not be further described herein.Numerous examples of level shifters exist throughout the prior art andmay be used without straying from the intended scope of the presentinvention. One such example of a level shifter is described in U.S. Pat.No. 5,214,602 to Lines. It should further be understood that while readand write command functions are not performed together, read and writemay be performed simultaneous with the refresh command function. Eachcommand insertion issues a “one” pulse with proper width that istrimmable. If this pulse width is too wide due to the process, voltageand/or temperature (PVT) variation of silicon, the pulse is merged tothe next pulse and generates a static low signal by internal logiccircuitry. In a further embodiment of the invention, the trimming of thepulse with of the control signal can be used to compensate for PVT.

The PMOS switch 210 shown in FIG. 2 advantageously increases thestability of the negative feedback amplifier. This switch 210 is onlyselectively active. During the power-up stage and the recovery of Vpplevel after re-enabling Vpp regulation by the Vpp-enable signal, theswitch 210 is turned on and operated together with the standby main PMOSdriver 211 to prevent the oscillation of Vpp from the active driver 212.A first active command such as read or write turns this switch 210 offso that the active driver 212 is independently operated.

The Miller compensation capacitor D4 shown in FIG. 2 prevents the Vpposcillation caused by the small load capacitance CLoad and high Vppcurrent. The steady driver can be operated with more stability by thisMiller capacitor that, for example, may be in the range of 10 pF. Asecond Miller compensation capacitor 221 is shown that has the samefunctionality as capacitor 220, though with a much reduced size, forexample, in the range of 2 pF. Reducing the size of this secondcapacitor 221 greatly facilitates application of the inventive VDCwithin smaller and smaller high-speed memory devices.

More specifically, elements 211 through 214 include four PMOS driversincluding a steady driver 211, an active driver 212, an active chargeinjector transistor 213 for read/write, and an active charge injectortransistor 214 for refresh. While only two active charge injectortransistors 213, 214 are shown, it should be understood that more thantwo are possible if additional command functions beyond read, write,read and refresh, write and refresh and refresh operations areperformed. Similarly, it should be readily apparent that only one activecharge injector transistor is required to perform the function inaccordance with the present invention if a refresh operation is notperformed in parallel to the read or write operations. For illustrationpurposes though, two charge injector transistors 213, 214 are shown.Such additional active charge injector transistors should therefore beconsidered to be well within the intended scope of the presentinvention. During power-up and all operation conditions, the steadydriver 211 takes control of Vpp. During power-up, the active driver 212has the same gate control level as the steady driver 211. However, afterasserting a first command, the active driver 212 operates independentlyaccording to Vpp level in order to respond quickly. The active chargeinjector transistors 213, 214 each operate in relation to acorresponding command insertion—e.g., the active charge injectortransistor 213 of read and write is activated upon a read command. Inthis way, two separate PMOS are able to easily react to the peak currentof single word line access and double word lines. One PMOS chargeinjection current is trimmed as much as the requirement of a singlecommand, that is, a single word line activation.

In practice, the steady driver 211 will have a large transistor widthrelative to the active driver 212. Moreover, each active charge injectortransistor 213, 214 will have even smaller transistor widths than theactive driver 212. As shown in FIG. 2, the active charge injectortransistors 213, 214 may have a small W/2 transistor width, but may be ¼width or less depending upon the given architecture and related load onthe given array. Such modifications are possible so long as the size ofthe transistors is enough to compensate for current load of performingan active cycle in the memory.

In operation, the active driver 212 moves down faster than the steadydriver 211 because of the small Miller compensation capacitor 220 andsmall driver size. However, drastic voltage drop is avoided by theactive charge injector transistor operation such that the active chargeinjector transistors 213, 214 preserve the Vpp level without usingcomplicated control logic decoding. It should be understood that thecharge injector transistor logic is implemented using delays and D-flipflops where the additional logic is negligible relative to known mannerof providing a huge reservoir capacitor to be able to keep Vpp at adesired level. Further, the fast response of the active charge injectortransistor elements 213, 214 does not require any feedback loop toensure the quick recovery of Vpp level. As well, PVT variation of theVpp level and current consumption can be tuned by pulse width controlfor process variation. Voltage and temperature variation are basicallyresolved with PMOS driver temperature and voltage variation of turn-oncurrent without trimming.

The three relevant states of operation in regard to the VDC of thepresent invention include power-up, standby mode, and consecutive activemode. During the power-up period, Vpp and control signals seek thecorresponding levels to follow the intermittent Vpp reference leveluntil a source voltage of Vpp (Vddq in this example) boosts up to thedetermined level. In this way, large current is not issued because of along power-up time. For standby mode, static DC current by transistorleakage is invoked where no significant transient from the Vpp loadcapacitor CLoad occurs and the steady and active drivers are tiedtogether through the PMOS switch 210. The regulator 230, 231 finds thefinal optimized Vpp level according to the Vpp reference voltage level.Any steady voltage drop of Vpp is compensated from the steady 211 andactive 212 drivers.

For consecutive active mode, huge and drastically sinking current fromthe Vpp load capacitor CLoad is demanded by the word line activation andrelevant circuitry such as the level shifter 251 or 261 into a word linecontrol block. Because of the slow response of the steady driver 211 andactive 212 driver at high frequency—e.g., 1 GHz operation of theconsecutive read or write with concurrent refresh that is a twice higherpeak current than single operation—, the Vpp level drops abruptly. Inorder to facilitate the recovery of this sudden drop of Vpp, the activecharge injector transistors are enabled depending on the command typessuch as single bank access (e.g., single read or write or refresh) anddouble bank access (e.g., read or write with concurrent refresh). Forsingle bank access, only one of two active charge injector transistors213, 214 are activated by a single pulse generated from a pulsegenerator (not shown) involved in the combinational logic block. Adouble bank access triggers both of the active charge injectortransistors 213, 214 so that the increased Vpp current is easilyrecovered.

From FIG. 3, power-up, standby, and consecutive active mode are showngraphically. While specific values are shown, they should be understoodto be only examples for purposes of illustration and not consideredlimiting. The consecutive active mode is of most concern, especiallywhere a command with concurrent refresh exists which has a double bankaccess invoking twice higher peak current from Vpp. In such instance,the quick response cannot be expected from the steady driver.Furthermore, the feedback provided to the comparator does not have asufficiently fast response in order to increase the current provided bythe active driver to restore the charge that the memory consumes duringan active mode. Without the use of the charge injector transistors, theVpp level would go down for example, less than 1.5v thus potentiallycreating problems for writing into the accessed cell. This is preventedby the use of the active charge injector transistor. When the activecharge injector transistor is enabled by its control signal, the losscharge of the Vpp output node is restored.

FIG. 4 is a simplified version of FIG. 2 that illustrates operation ofthe VDC circuit in power-up and standby modes. FIG. 4 shows that thesteady 211 and active 212 drivers are driven by the two regulators 230,231 so as to determine the Vpp level for power-up and standby mode. Theregulator 230, 231 takes a control of the Vpp level with Vpp ref that isset by the reference generator. During power-up and standby mode (i.e.,steady state), only the steady 211 and active 212 drivers are activatedand electrically tied together through PMOS switch 210. However, asfurther illustrated by FIG. 5, all drivers 211 through 214 including theactive charge injector transistors 213, 214 are activated to quicklyrestore the Vpp level without any significant drop of Vpp level.Further, the pulse width of the control signal provided to the gates ofthe charge injector transistors 213, 214 is trimmable so that the VDC200 b can supply as much as whatever Vpp current is sunk by the activeoperations of the memory.

FIG. 6 is a graphical illustration that shows a simplified version ofVpp with an addition of an active charge injector transistor inaccordance with the present invention and Vpp without an active chargeinjector transistor. The top graph in FIG. 6 shows a load current withtwo peaks in the load that indicate an active cycle in the memory arrayin which an operation occurs such as read, write, or refresh. The impacton Vpp without the benefit of an active charge injector transistor isshown at the bottom graph where Vpp initially drops upon increase inload current, fails to fully recover, drops again upon a second increasein load current, and subsequently slowly recovers to full value. Thegraph labelled “Vpp with charge injector transistor” shows the impact ofthe present invention on Vpp whereby Vpp suffers from only minorfluctuations and regains full value quickly after each increase in loadcurrent. As the active charge injector transistor in accordance with thepresent invention is digitally controlled, the graph labelled “chargeinjector transistor control signal” is provided to indicate activationof the active charge injector transistor. Though over-simplified forpurposes of illustration, it should be noted that a slight delay ofcourse occurs from the time activity in the array begins until the timethat a pulse is provided to the active charge injector transistor. In afurther embodiment of the invention, the activation of the chargeinjector transistor can be set to start at the time that the word linedrivers are activated, so the current load is quickly compensated. Inyet another embodiment of the invention, the activation of the chargeinjector transistor can be set to be earlier than the activation of theword line drivers so a drop on the Vpp voltage level is avoidedcompletely. It would be obvious to someone skilled in the art thattrimming can be used to set the best results and achieve the beststability of the Vpp voltage or to better accommodate the particularneeds of their memories.

The present invention is useful in any small sized memory withhigh-speed operation including, but not limited to, embedded DRAM. It isespecially useful where developers are unable to acquire enough spaceon-chip to place a huge reservoir capacitor, especially within deviceshaving an operating frequency over 700 MHz. For large arrays, thepresent invention may be provided in a multi-bank version such thatseveral separate banks of memory on a single chip may be served bymultiple VDCs in accordance with the present invention. For instance,four memory blocks may be made independent such that access can differwhere perhaps a first memory block may be read while a second memoryblock may be written. In such instance, the present invention may beprovided in a multiple format to serve each block independently. Thepresent invention also reduces the need for embedded memory to have anindependent external power supply to resolve large drops within anyinternal power source. The present invention further presents a VDC withfast operation of over 1 GHz with a large Vpp current and a very smallphysical size requirement that are all important features of embeddedmemory products.

The above-described embodiments of the present invention are intended tobe examples only. Alterations, modifications and variations may beeffected to the particular embodiments by those of skill in the artwithout departing from the scope of the invention, which is definedsolely by the claims appended hereto.

1. A method for supplying a voltage to a random access memory circuitwithin an integrated circuit device, the integrated circuit devicehaving an input pin for receiving an external high voltage, the methodcomprising: (a) connecting the input pin through a series passtransistor to a Vpp output node; and (b) regulating said series passtransistor to generate a substantially constant high voltage Vpp that isgreater than the logic “1” voltage level stored in a memory cell.